Overcurrent protection circuit

ABSTRACT

In order both to accommodate instantaneous current as well as overcurrent protection in accordance with the load, an overcurrent protection circuit has: a threshold value generation unit that, in accordance with a threshold value control signal, switches between setting an overcurrent detection threshold value to a first set value (∝ Iref) and a second set value (∝ Iset) lower than the first set value; an overcurrent detection unit that compares a sense signal in accordance with the current being monitored and the overcurrent detection value and generates an overcurrent protection signal; a reference value generation unit that generates a reference value (∝ Iset) in accordance with the seconds set value; a comparison unit that compares the sense signal and the reference value, and generates a comparison signal; and a threshold value control unit that monitors the comparison signal, and generates a threshold value control signal.

TECHNICAL FIELD

The present invention relates to an overcurrent protection circuit.

BACKGROUND ART

Conventionally, many of semiconductor integrated circuit devices areprovided with an overcurrent protection circuit as one of theirabnormality protection circuits. For example, an in-vehicle IPD(intelligent power device) is provided with an overcurrent protectioncircuit, which restricts the amount of output current flowing through apower transistor not to exceed an overcurrent set value, for the purposeof preventing the device from breaking in a case of a short-circuit in aload connected to the power transistor. In recent years, there has beenproposed an overcurrent protection circuit that is capable of adjustingan overcurrent set value as necessary by using an external resistor.

Examples of conventional technologies related to the above are disclosedin Patent Document 1 and Patent Document 2 listed below.

CITATION LIST Patent Literature

Patent Document 1: Japanese Patent Application Publication No.2015-46954

Patent Document 2: Japanese Patent Application Publication No.2012-211805

SUMMARY OF INVENTION Technical Problem

However, loads connected to a power transistor include one (such as acapacitive load) that requires a large output current to instantaneouslyflow therethrough in its normal operation. In a case where a target tobe monitored is such an output current, with conventional overcurrentprotection circuits, having a single overcurrent set value, it isdifficult to achieve both the securing of the instantaneous current andovercurrent protection suitable for the load.

In particular, in recent years, in-vehicle ICs are required to complywith ISO26262 (the international standard for the functional safety ofelectrical electronics systems in production automobiles), and as toin-vehicle IPDs, too, a higher reliability design has become important.

The invention disclosed herein has been made in view of theabove-mentioned problem found by the inventors of the present invention,and an object thereof is to provide an overcurrent protection circuitcapable of achieving both the securing of an instantaneous current andovercurrent protection suitable for a load.

Solution to Problem

An overcurrent protection circuit disclosed herein includes a thresholdgenerator which switches, in accordance with a threshold control signal,whether an overcurrent detection threshold should be a first set valueor a second set value which is lower than the first set value, anovercurrent detector which compares a sense signal in accordance with amonitored current with the overcurrent detection threshold and therebygenerates an overcurrent protection signal, a reference value generatorwhich generates a reference value in accordance with the second setvalue, a comparison section which compares the sense signal with thereference value and thereby generates a comparison signal, and athreshold controller which monitors the comparison signal and therebygenerates the threshold control signal (first configuration).

In the overcurrent protection circuit having the first configuration,when the overcurrent detection threshold has been set to the first setvalue, the threshold controller may generate the threshold controlsignal such that the overcurrent detection threshold is switched to thesecond set value at a time point when a mask period elapses with thesense signal maintained above the reference value (secondconfiguration).

In the overcurrent protection circuit having the second configuration,when the overcurrent detection threshold has been set to the second setvalue, the threshold controller may generate the threshold controlsignal such that the overcurrent detection threshold is switched to thefirst set value at a time point when the sense signal falls below thereference value (third configuration).

In the overcurrent protection circuit having the second or thirdconfiguration, the mask period may be a variable value (fourthconfiguration).

In the overcurrent protection circuit having any one of the first tofourth configurations, the first set value may be a fixed value, and thesecond set value may be a variable value (fifth configuration).

A semiconductor integrated circuit device disclosed herein includes,integrated therein, a power transistor which switches a current path,through which an output current flows, between a conducting state and acutoff state, an output current monitor which generates a sense signalin accordance with the output current, a gate controller which generatesa driving signal for the power transistor in accordance with a controlsignal, and the overcurrent protection circuit having any one of thefirst to fifth configurations which monitors the sense signal andthereby generates an overcurrent protection signal. Here, the gatecontroller is provided with a function of forcibly turning off the powertransistor in accordance with the overcurrent protection signal (sixthconfiguration).

The semiconductor integrated circuit device having the sixthconfiguration may further include, integrated therein, a signal outputsection which selectively outputs, to outside the device, one of adetection result of the output current and an abnormality flag as astatus notification signal (seventh configuration).

An electronic apparatus disclosed herein includes the semiconductorintegrated circuit device having the sixth or seventh configuration, anda load connected to the semiconductor integrated circuit device (eighthconfiguration).

In the electronic apparatus having the eighth configuration, the loadmay be a bulb lamp, a relay coil, a solenoid, a light emitting diode, ora motor (ninth configuration).

A vehicle disclosed herein includes the electronic apparatus having theeighth or ninth configuration (tenth configuration).

An overcurrent protection circuit disclosed herein includes a firstthreshold generator which switches, in accordance with a first thresholdcontrol signal, whether a first overcurrent detection threshold shouldbe a first set value or a second set value which is lower than the firstset value, a second threshold generator which switches, in accordancewith a second threshold control signal, whether a second overcurrentdetection threshold should be a third set value or a fourth set valuewhich is lower than the third set value, a first overcurrent detectorwhich compares a first sense signal in accordance with a first monitoredcurrent with the first overcurrent detection threshold and therebygenerates a first overcurrent protection signal, a second overcurrentdetector which compares a second sense signal in accordance with asecond monitored current with the second overcurrent detection thresholdand thereby generates a second overcurrent protection signal, a firstreference value generator which generates a first reference value inaccordance with the second set value, a second reference value generatorwhich generates a second reference value in accordance with the fourthset value, a first comparison section which compares the first sensesignal with the first reference value and thereby generates a firstcomparison signal, a second comparison section which compares the secondsense signal with the second reference value and thereby generates asecond comparison signal, and a threshold controller which monitors boththe first comparison signal and the second comparison signal and therebygenerates the first threshold control signal and the second thresholdcontrol signal (eleventh configuration).

In the overcurrent protection circuit having the eleventh configuration,the threshold controller may include an external terminal for externallyconnecting a capacitor, a comparator which compares a charge voltagewhich appears at the external terminal with a predetermined referencevoltage and thereby generates an internal signal, a first flip-flopwhich generates the first threshold control signal in accordance withthe internal signal and the first comparison signal, a second flip-flopwhich generates the second threshold control signal in accordance withthe internal signal and the second comparison signal, a dischargecontroller which performs discharge control of the capacitor inaccordance with the internal signal, and a charge controller whichperforms charge control of the capacitor in accordance with both thefirst comparison signal and the second comparison signal (twelfthconfiguration).

In the overcurrent protection circuit having the twelfth configuration,the discharge controller may accept input of not only the internalsignal but also the first comparison signal, the second comparisonsignal, the first threshold control signal, and the second thresholdcontrol signal, and in a case where, after a logic-level change occursin one of the first comparison signal and the second comparison signaland a charging operation of the capacitor is started, a logic-levelchange occurs in an other of the first comparison signal and the secondcomparison signal before the charge voltage becomes higher than thereference voltage, the capacitor may be discharged (thirteenthconfiguration).

In the overcurrent protection circuit having the thirteenthconfiguration, the threshold controller may further include a firstdelay section which gives a delay to the first comparison signal andthereby generates a first delay signal, and a second delay section whichgives a delay to the second comparison signal and thereby generates asecond delay signal, and the first delay signal and the second delaysignal, instead of the first comparison signal and the second comparisonsignal, may be inputted to the first flip-flop and the second flip-flop,respectively (fourteenth configuration).

In the overcurrent protection circuit having any one of the eleventh tofourteenth configurations, the first set value and the third set valuemay each be a fixed value, and the second set value and the fourth setvalue may each be a variable value (fifteenth configuration).

A semiconductor integrated circuit device disclosed herein includes,integrated therein, a first power transistor which switches a firstcurrent path, through which a first output current flows, between aconducting state and a cutoff state, a second power transistor whichswitches a second current path, through which a second output currentflows, between a conducting state and a cutoff state, a first outputcurrent monitor which generates a first sense signal in accordance withthe first output current, a second output current monitor whichgenerates a second sense signal in accordance with the second outputcurrent, a first gate controller which generates a first driving signalfor the first power transistor in accordance with a first controlsignal, a second gate controller which generates a second driving signalfor the second power transistor in accordance with a second controlsignal, and the overcurrent protection circuit having any one of theeleventh to fifteenth configurations which monitors the first sensesignal and the second sense signal and thereby generates a firstovercurrent protection signal and a second overcurrent protectionsignal. Here, the first gate controller and the second gate controllerhave functions of forcibly turning off the first power transistor andthe second power transistor in accordance with the first overcurrentprotection signal and the second overcurrent protection signal,respectively (sixteenth configuration).

The semiconductor integrated circuit device having the sixteenthconfiguration may further include, integrated therein, a first signaloutput section which generates one of a detection result of the firstoutput current and an abnormality flag as a first status notificationsignal, a second signal output section which generates one of adetection result of the second output current and an abnormality flag asa second status notification signal, and a multiplexer which selectivelyoutputs one of the first status notification signal and the secondstatus notification signal to outside the device (seventeenthconfiguration).

An electronic apparatus disclosed herein includes the semiconductorintegrated circuit device having the sixteenth or seventeenthconfiguration, a first load connected to the first power transistor, anda second load connected to the second power transistor (eighteenthconfiguration).

In the electronic apparatus having the eighteenth configuration, thefirst load and the second load may each be a bulb lamp, a relay coil, asolenoid, a light emitting diode, or a motor (nineteenth configuration).

A vehicle disclosed herein includes the electronic apparatus having theeighteenth or nineteenth configuration (twentieth configuration).

Advantageous Effects of Invention

According to the invention disclosed herein, it is possible to providean overcurrent protection circuit capable of achieving both the securingof an instantaneous current and overcurrent protection suitable for aload.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating a first embodiment of asemiconductor integrated circuit device;

FIG. 2 is a block diagram illustrating an example of a configuration ofa signal output section;

FIG. 3 is a block diagram illustrating an example of a configuration ofa gate controller;

FIG. 4 is a block diagram illustrating an example of a configuration ofan overcurrent protection circuit;

FIG. 5 is a circuit diagram illustrating an example of a configurationof a first current generator;

FIG. 6 is a circuit diagram illustrating an example of a configurationof a second current generator;

FIG. 7 is a circuit diagram illustrating an example of a configurationof a threshold voltage generator and an overcurrent detector;

FIG. 8 is a schematic diagram illustrating an example of an overcurrentset value;

FIG. 9 is a circuit diagram illustrating an example of configurations ofa reference voltage generator and a comparison section;

FIG. 10 is a circuit diagram illustrating an example of a configurationof a threshold controller;

FIG. 11 is a timing chart illustrating an example of an overcurrentprotection operation:

FIG. 12 is a flowchart illustrating an example of a threshold switchingoperation;

FIG. 13 is a schematic diagram illustrating a first usage example of anovercurrent protection circuit;

FIG. 14 is a schematic diagram illustrating a second usage example ofthe overcurrent protection circuit;

FIG. 15 is a block diagram illustrating a second embodiment of thesemiconductor integrated circuit device;

FIG. 16 is a block diagram illustrating an example of a configuration ofa two-channel overcurrent protection circuit;

FIG. 17 is a block diagram illustrating a first example of a thresholdcontroller;

FIG. 18 is a timing chart illustrating a threshold switching operationof the first example;

FIG. 19 is a timing chart illustrating a disadvantage of the firstexample;

FIG. 20 is a block diagram illustrating a second example of thethreshold controller;

FIG. 21 is a block diagram illustrating an example of a configuration ofa discharge controller;

FIG. 22 is a timing chart illustrating a threshold switching operationof the second example;

FIG. 23 is a timing chart illustrating a disadvantage of the secondexample;

FIG. 24 is a block diagram illustrating a third example of the thresholdcontroller;

FIG. 25 is a timing chart illustrating a threshold switching operationof the third example;

FIG. 26 is a flowchart illustrating an example of a threshold switchingoperation;

FIG. 27 is a block diagram illustrating an example where a multiplexeris introduced; and

FIG. 28 is an external view of a vehicle, illustrating an example of aconfiguration of the vehicle.

DESCRIPTION OF EMBODIMENTS Semiconductor Integrated Circuit Device(First Embodiment)

FIG. 1 is a block diagram illustrating a first embodiment of asemiconductor integrated circuit device. The semiconductor integratedcircuit device 1 of the present embodiment is an in-vehicle high-sideswitch IC (a kind of in-vehicle IPD) which achieves a conducting/cutoffstate between an application terminal of a power supply voltage VBB anda load 3 in accordance with an instruction from an ECU (electroniccontrol unit) 2.

Here, the semiconductor integrated circuit device 1 includes externalterminals T1 to T4 as means for establishing electrical connection withoutside the device. The external terminal T1 is a power supply terminal(a VBB pin) for receiving supply of power supply voltage VBB (12V, forexample) from an unillustrated battery. The external terminal T2 is aload connection terminal (an OUT pin) for externally connecting a load 3(such as a bulb lamp, a relay coil, a solenoid, a light emitting diode,or a motor). The external terminal T3 is a signal input terminal (an INpin) for accepting external input of an external control signal Si fromthe ECU 2. The external terminal T4 is a signal output terminal (a SENSEpin) for externally outputting a status notification signal So to theECU 2. Here, between the external terminal T4 and a ground terminal, anexternal sense resistor 4 is externally connected.

The semiconductor integrated circuit device 1 includes, integratedtherein, an NMOSFET 10, an output current monitor 20, a gate controller30, a control logic section 40, a signal input section 50, an internalpower supply 60, an abnormality protection section 70, an output currentdetector 80, and a signal output section 90.

The NMOSFET 10 is a high-withstanding-voltage (withstanding voltage of42 V, for example) power transistor having a drain connected to theexternal terminal T1 and having a source connected to the externalterminal T2. The NMOSFET 10 connected in this fashion functions as aswitch element (a high-side switch) for switching a current path fromthe application terminal of the power supply voltage VBB to the groundterminal via the load 3 between a conducting state and a cutoff state.The NMOSFET 10 is turned on when a gate driving signal G1 is at highlevel, and is turned off when the gate driving signal G1 is at lowlevel.

The NMOSFET 10 may be designed such that its on-resistance value isseveral tens of mΩ. However, as the on-resistance value of the NMOSFET10 is lower, it is more likely that an overcurrent flows at a time ofground fault of the external terminal T2 (i.e., at a time ofshort-circuit to the ground terminal or to a terminal with an equivalentlow potential), which may result in abnormal heat generation.Accordingly, as the on-resistance value of the NMOSFET 10 is lower, anovercurrent protection circuit 71 and a temperature protection circuit73, of which both will be described later, become more important.

The output current monitor 20 includes NMOSFETs 21 and 21′ and a senseresistor 22, and generates a sense voltage Vs (=corresponding to a sensesignal) in accordance with an output current Io flowing through theNMOSFET 10.

Both of the NMOSFETs 21 and 21′, which are mirror transistors connectedin parallel to the NMOSFET 10, respectively generate sense currents Isand Is′ in accordance with the output current Io. The size ratio betweenthe NMOSFET 10 and the NMOSFETs 21 and 21′ is m:1 (where m>1).Accordingly, the sense currents Is and Is′ are equal to 1/m of theoutput current Io. Like the NMOSFET 10, the NMOSFETs 21 and 21′ areturned on when a gate driving signal G1 is at high level, and are turnedoff when a gate voltage G2 is at low level.

The sense resistor 22 (resistance: Rs) is connected between a source ofthe NMOSFET 21 and the external terminal T2, and is a current/voltageconversion element that generates a sense voltage Vs (=Is×Rs+Vo, whereVo represents an output voltage appearing at the external terminal T2)in accordance with the sense current Is.

The gate controller 30 generates the gate driving signal G1 withincreased current capability of the gate control signal S1, and outputsthe gate driving signal G1 to gates of the NMOSFETs 10 and 21, therebycontrolling the turning-on/off of the NMOSFETs 10 and 21. Here, the gatecontroller 30 includes a function of forcibly turning off the NMOSFETs10 and 21 independent of the gate control signal S1 in a case where anovercurrent protection signal S71 is at a logic level of a time whenabnormality is detected.

The control logic section 40, on receiving an internal power supplyvoltage Vreg, generates the gate control signal S1. For example, whenthe external control signal S1 is at high level (=a logic level forturning on the NMOSFET 10), the internal power supply voltage Vreg issupplied from the internal power supply 60, such that the control logicsection 40 is brought into an operating state, and the gate controlsignal S1 becomes high level (=Vreg). On the other hand, when theexternal control signal Si is at low level (=a logic level for turningoff the NMOSFET 10), the internal power supply voltage Vreg is notsupplied from the internal power supply 60, such that the control logicsection 40 is brought into a non-operating state, and the gate controlsignal S1 becomes low level (=GND). In addition, the control logicsection 40 monitors various abnormality protection signals (theovercurrent protection signal S71, an open protection signal S72, atemperature protection signal S73, and a voltage reduction protectionsignal S74). The control logic section 40 also has a function ofgenerating an output switching signal S2 in accordance with results ofthe monitoring of the overcurrent protection signal S71, the openprotection signal S72, and the temperature protection signal S73 amongthe above-mentioned abnormality protection signals.

The signal input section 50 is a Schmitt trigger, which receives theexternal control signal Si from the external terminal T3, and transmitsit to the control logic section 40 and the internal power supply 60.Here, the external control signal Si becomes high level to turn on theNMOSFET 10, and becomes low level to turn off the NMOSFET 10.

The internal power supply 60 generates a predetermined internal powersupply voltage Vreg from the power supply voltage VBB, and supplies itto each of the sections of the semiconductor integrated circuit device1. Here, whether or not the internal power supply 60 is operable iscontrolled in accordance with the external control signal Si. Morespecifically, the internal power supply 60 is in an operating state whenthe external control signal Si is at high level, and is in anon-operating state when the external control signal Si is at low level.

The abnormality protection section 70 is a circuit block that detects avariety of abnormalities occurring in the semiconductor integratedcircuit device 1, and includes an overcurrent protection circuit 71, anopen protection circuit 72, a temperature protection circuit 73, and avoltage reduction protection circuit 74.

The overcurrent protection circuit 71 generates the overcurrentprotection signal S71 in accordance with a result of monitoring of thesense voltage Vs (whether or not an overcurrent abnormality of theoutput current Io has occurred). Here, the overcurrent protection signalS71 becomes low level when no abnormality is detected, and becomes highlevel when an abnormality is detected, for example.

The open protection circuit 72 generates the open protection signal S72in accordance with a result of monitoring of the output voltage Vo(whether or not an open abnormality of the load 3 has occurred). Here,for example, the open protection signal S72 becomes low level when noabnormality is detected, and becomes high level when an abnormality isdetected.

The temperature protection circuit 73 includes a temperature detectionelement (not shown) for detecting abnormal heat generation in thesemiconductor integrated circuit device 1 (in particular, around theNMOSFET 10), and generates the temperature protection signal S73 inaccordance with a result of the detection (whether or not abnormal heatgeneration has occurred). Here, for example, the temperature protectionsignal S73 becomes low level when no abnormality is detected, andbecomes high level when an abnormality is detected.

The voltage reduction protection circuit 74 generates the voltagereduction protection signal S74 in accordance with a result ofmonitoring of the power supply voltage VBB or the internal power supplyvoltage Vreg (whether or not a reduced voltage abnormality hasoccurred). Here, for example, the voltage reduction protection signalS74 becomes low level when no abnormality is detected, and becomes highlevel when an abnormality is detected.

The output current detector 80 makes a source voltage of the NMOSFET 21′equal to the output voltage Vo using bias means (not shown), and therebygenerates a sense current Is′ (=Io/m) in accordance with the outputcurrent Io, and outputs it to the signal output section 90.

Based on an output selection signal S2, the signal output section 90selectively outputs one of the sense current Is′ (=corresponding to thedetection result of the output current Io) and a fixed voltage V90(=corresponding to an abnormality flag, which is not clearly illustratedin the figure) to the external terminal T4. Here, in a case where thesense current Is′ is selectively outputted, an output detection voltageV80 (=Is′×R4) obtained by current/voltage conversion of the sensecurrent Is′ by means of the external sense resistor 4 (resistance: R4)is transmitted as the status notification signal So to the ECU 2. Here,the output detection voltage V80 increases with the output current Io,and decreases with the output current Io. On the other hand, when thefixed voltage V90 is selectively outputted, the fixed voltage V90 istransmitted as the status notification signal So to the ECU 2.

<Signal Output Section>

FIG. 2 is a block diagram illustrating an example of a configuration ofthe signal output section 90. The signal output section 90 of thisconfiguration example includes a selector 91. The selector 91selectively outputs the sense current Is′ to the external terminal T4when the output selection signal S2 is at a logic level (for example,low level) of when no abnormality is detected, and, when the outputselection signal S2 is at a logic level (for example, high level) ofwhen abnormality is detected, the selector 91 selectively outputs thefixed voltage V90 to the external terminal T4. Here, the fixed voltageV90 is set to a voltage value higher than an upper limit value of theabove-mentioned output detection voltage V80.

According to such a signal output section 90, it is possible to transmitboth the detection result of the output current Io and the abnormalityflag to the ECU 2 by using a single status notification signal So, andthus to contribute to a reduced number of external terminals. Here, in acase of reading a current value of the output current Io from the statusnotification signal So, an A/D (analog-to-digital) conversion isperformed on the status signal So. On the other hand, in a case ofreading an abnormality flag from the status signal So, the logic levelof the status signal So is determined by using a threshold that isslightly lower than the fixed voltage V90.

<Gate Controller>

FIG. 3 is a block diagram illustrating an example of a configuration ofthe gate controller 30. The gate controller 30 of this configurationexample includes a gate driver 31, an oscillator 32, a charge pump 33, adamper 34, and an NMOSFET 35.

The gate driver 31 is connected between an output terminal of the chargepump 33 (=an application terminal of a boosted voltage VG) and theexternal terminal T2 (an application terminal of the output voltage Vo),and generates the gate driving signal G1 with increased currentcapability of the gate control signal S1. Here, the gate driving signalG1 is at high level (=VG) when the gate control signal S1 is at highlevel, and is at low level (=Vo) when the gate control signal S1 is atlow level.

The oscillator 32 generates a clock signal CLK of a predeterminedfrequency, and outputs it to the charge pump 33. Whether or not theoscillator 32 is operable is controlled according to an enable signal Sareceived from the control logic section 40.

The charge pump 33 generates the boosted voltage VG, which is higherthan the power supply voltage VBB, by driving a flying capacitor byusing the clock signal CLK. Here, whether or not the charge pump 33 isoperable is controlled in accordance with an enable signal Sb receivedfrom the control logic section 40.

The damper 34 is connected between the external terminal T1 (theapplication terminal of the power supply voltage VBB) and the gate ofthe NMOSFET 10. In an application in which an inductive load 3 isconnected to the external terminal T2, the output voltage Vo is causedto be a negative voltage (<GND) by a counter electromotive force of theload 3 when switching the NMOSFET 10 from ON to OFF. Thus, the damper 34(what is called an active clamp circuit) is provided for energyabsorption.

A drain of the NMOSFET 35 is connected to the gate of the NMOSFET 10. Asource of the NMOSFET 35 is connected to the external terminal T2. Agate of the NMOSFET 35 is connected to an application terminal of theovercurrent protection signal S71.

In the gate controller 30 of the present configuration example, theNMOSFET 35 is turned off when the first overcurrent protection signalS71 is at low level (=a logic level of when no abnormality is detected),and accordingly the gate driving signal G1 is applied to the NMOSFET 10in a normal fashion. On the other hand, when the overcurrent protectionsignal S71 is at high level (=a logic level of when abnormality isdetected), the NMOSFET 35 is turned on, and accordingly a short circuitis caused between the gate and the source of the NMOSFET 10.

In this fashion, the gate controller 30 of the present configurationexample has a function of controlling the gate driving signal G1 suchthat the NMOSFET 10 is forcibly turned off when the overcurrentprotection signal S71 is at high level (=the logic level of whenabnormality is detected).

<Overcurrent Protection Circuit>

FIG. 4 is a block diagram illustrating an example of a configuration ofthe overcurrent protection circuit 71. The overcurrent protectioncircuit 71 of the present configuration example includes a first currentgenerator 110, a second current generator 120, a threshold voltagegenerator 130, an overcurrent detector 140, a reference voltagegenerator 150, a comparison section 160, and a threshold controller 170.

The first current generator 110 generates a first current Iref, andoutputs it to the threshold voltage generator 130. The first currentIref has a current value that is fixed inside the semiconductorintegrated circuit device 1.

The second current generator 120 generates a second current Iset, andoutputs it to the threshold voltage generator 130. The second currentIset has a current value that is adjustable as necessary from outsidethe semiconductor integrated circuit device 1.

The threshold voltage generator 130 switches, in accordance with athreshold control signal S170, whether to set a threshold voltage Vth(=corresponding to an overcurrent detection threshold) to an internalset value VthH or to an external set value VthL (VthH>VthL). Here, theinternal set value VthH is a fixed value (=corresponding to a first setvalue) which is set in accordance with the first current Iref. On theother hand, the external set value VthL is a variable value(=corresponding to a second set value) which is set in accordance withthe second current Iset.

The overcurrent detector 140 compares the sense voltage Vs with thethreshold voltage Vth, and thereby generates the overcurrent protectionsignal S71.

The reference voltage generator 150 generates a reference voltage VIset(=corresponding to a reference value) in accordance with the secondcurrent Iset.

The comparison section 160 compares the sense voltage Vs with thereference voltage VIset, and thereby generates a comparison signal VCMP.

The threshold controller 170 monitors the comparison signal VCMP, andthereby generates the threshold control signal S170. Here, the thresholdcontrol signal S170 becomes low level when the internal set value VthHshould be selected as the threshold voltage Vth, and becomes high levelwhen the external set value VthL should be selected as the thresholdvoltage Vth.

<First Current Generator>

FIG. 5 is a circuit diagram illustrating an example of a configurationof the first current generator 110. The first current generator 110 ofthe present configuration example includes an operational amplifier 111,an NMOSFET 112, and a resistor 113 (resistance: R113).

A power supply terminal of the operational amplifier 111 is connected toan application terminal of the internal power supply voltage Vreg. Areference potential terminal of the operational amplifier 111 isconnected to a ground terminal GND. A non-inverting input terminal (+)of the operational amplifier 111 is connected to an application terminalof a reference voltage Vref (for example, a band-gap reference voltagewhich is unlikely to be affected by power supply variation, temperaturevariation, etc.). An inverting input terminal (−) of the operationalamplifier 111 and a source of the NMOSFET 112 are connected to a firstterminal of the resistor 113. A second terminal of the resistor 113 isconnected to the ground terminal GND. An output terminal of theoperational amplifier 111 is connected to a gate of the NMOSFET 112. Adrain of the NMOSFET 112 is connected to an output terminal of the firstcurrent Iref.

The operational amplifier 111 connected as described above controls thegate of the transistor 112 such that the non-inverting input terminal(+) and the inverting input terminal (−) are imaginarilyshort-circuited. As a result, the first current Iref (=Vref×R113) havinga fixed value flows through the resistor 113.

<Second Current Generator>

FIG. 6 is a circuit diagram illustrating an example of a configurationof the second current generator 120. The second current generator 120 ofthe present configuration example includes an operational amplifier 121,an NMOSFET 122, a resistor 123 (resistance: R123), and an externalterminal SET.

To a power supply terminal of the operational amplifier 121, theapplication terminal of the internal power supply voltage Vreg isconnected. A reference potential terminal of the operational amplifier121 is connected to the ground terminal GND. A non-inverting inputterminal (+) of the operational amplifier 121 is connected to anapplication terminal of the reference voltage Vref. An inverting inputterminal (−) of the operational amplifier 121 and a source of theNMOSFET 122 are connected to the external terminal SET. An outputterminal of the operational amplifier 121 is connected to a gate of theNMOSFET 122. A drain of the NMOSFET 122 is connected to an outputterminal of the second current Iset. The resistor 123 is, outside thesemiconductor integrated circuit device 1, connected between theexternal terminal SET and the ground terminal GND.

The operational amplifier 121 connected as described above controls thegate of the transistor 122 such that the non-inverting input terminal(+) and the inverting input terminal (−) are imaginarilyshort-circuited. As a result, the second current Iset (=Vref×R123) inaccordance with the resistance R123 of the resistor 123 flows throughthe resistor 123. That is, the second current Iset increases with theresistance R123, and decreases with the resistance R123. Thus, it ispossible to adjust the second current Iset as necessary by using theresistor 123, which is connected externally. Here, by employing acascode circuit in a differential stage in the operational amplifier121, it is possible to set the second current Iset more accurately.

<Threshold Voltage Generator, Overcurrent Detector>

FIG. 7 is a circuit diagram illustrating an example of a configurationof the threshold voltage generator 130 and the overcurrent detector 140.The threshold voltage generator 130 includes a current source 131, aresistor 132, and a current mirror 133. On the other hand, theovercurrent detector 140 includes a comparator 141.

The current source 131 is connected between a current input terminal ofthe current mirror 133 and an application terminal of a constant voltageVBBM5, and selectively outputs one of the first current Iref and thesecond current Iset in accordance with the threshold control signalS170. More specifically, the current source 131 selectively outputs thefirst current Iref when the threshold control signal S170 is at lowlevel, and selectively outputs the second current Iset when thethreshold control signal S170 is at high level.

The resistor 132 is connected between a current output terminal of thecurrent mirror 133 and the application terminal of the output voltage Vo(that is, the external terminal T2), and the resistance value of theresistor 132 is switched to one of a first resistance Rref1 and a secondresistance Rref2 in accordance with the threshold control signal S170.More specifically, the resistance of the resistor 132 is the firstresistance Rref1 when the threshold control signal S170 is at low level,and is the second resistance Rref2 when the threshold control signalS170 is at high level.

The current mirror 133 operates on receiving a supply of a constantvoltage VBB_REF and the boosted voltage VG, mirrors the first currentIref or the second current Iset inputted thereto from the current source131, and outputs it to the resistor 132. Thus, at the current outputterminal of the current mirror 133 (a high potential terminal of theresistor 132), the threshold voltage Vth is generated of which thevoltage value is switched in accordance with the threshold controlsignal S170. More specifically, the threshold voltage Vth is theinternal set value VthH (=Iref×Rref1) when the threshold control signalS170 is at low level, and the threshold voltage Vth is the external setvalue VthL (=Iset×Rref2) when the threshold control signal S170 is athigh level. Here, the current mirror 133 functions also as a levelshifter which delivers the first current Iref or the second current Isetfrom a first power supply system (a VBB_REF−VBBM5 system) to a secondpower supply system (a VG-Vo system).

Here, the constant voltage VBB_REF and the constant voltage VBBM5 areboth reference voltages generated in the semiconductor integratedcircuit device 1, and, for example, VBB_REF≈VBB and VBBM5≈VBBSV.

A power supply terminal of the comparator 141 is connected to theapplication terminal of the boosted voltage VG. A reference potentialterminal of the comparator 141 is connected to the application terminalof the output voltage Vo (the external terminal T2). A non-invertinginput terminal (+) of the comparator 141 is connected to an applicationterminal of the sense voltage Vs. An inverting input terminal (−) of thecomparator 141 is connected to an application terminal of the thresholdvoltage Vth. The comparator 141 connected in this fashion compares thesense voltage Vs with the threshold voltage Vth, and thereby generatesthe overcurrent protection signal S71. The overcurrent protection signalS71 becomes low level (a logic level of when an overcurrent is detected)when the sense voltage Vs is lower than the threshold voltage Vth, andbecomes high level (a logic level of when an overcurrent is notdetected) when the sense voltage Vs is higher than the threshold voltageVth.

FIG. 8 is a schematic diagram illustrating an example of an overcurrentset value. As already described above, the threshold voltage Vth, whichis compared with the sense voltage Vs, is switched to one of theinternal set value VthH and the external set value VthL in accordancewith the threshold control signal S170. This is equivalent to that theovercurrent set value Iocp, which is compared with the output currentIo, is switched to one of the internal set value IocpH and the externalset value IocpL.

Here, the internal set value IocpH is desirably be a fixed value (forexample, about 15 A) in accordance with an on-resistance, a devicewithstanding voltage, or the like of the NMOSFET 10 such that thesemiconductor integrated circuit device 1 will not be destroyed even ina case where short-circuit abnormality has occurred in the load 3. Thus,the internal set value IocpH is provided for a dedicated purpose ofprotecting the semiconductor integrated circuit device 1 itself, andoften greatly deviates from a steady value of the output current Io.

On the other hand, in view of that an abnormal value of the outputcurrent Io depends on the load 3, the external set value IocpL isdesirably a variable value (for example, 1 A to 10 A) in accordance withthe load 3. For example, when a bulb lamp is driven, the output currentIo is generally greater than when a solenoid is driven. In view of this,when a bulb lamp is driven, the external set value IocpL should be sethigher than when a solenoid is driven. The output current Io when alight emitting diode is driven is generally smaller than when a solenoidis driven. In view of this, when a light emitting diode is driven, theexternal set value IocpL should be set lower than when a solenoid isdriven.

Now, the load 3 as a target to be driven by the semiconductor integratedcircuit device 1 can be a load that requires a large output current Ioto instantaneously flow therethrough in its normal operation. Forexample, at a time of activating a bulb lamp, a rush current that islarger than in a steady-state operation instantaneously flows in thebulb lamp. Depending on the load 3, the output current Io at the time ofactivating the load 3 may be different, by several tens of times, fromthat of the time of the steady-state operation.

Thus, to achieve both the securing of an instantaneous current andovercurrent protection suitable for the load 3, it is necessary toswitch the overcurrent set value Iocp, which is compared with the outputcurrent Io (and thus the threshold voltage Vth, which is compared withthe sense voltage Vs) with appropriate timing.

Hereinafter, detailed descriptions will be given of means (the referencevoltage generator 150, the comparison section 160, and the thresholdcontroller 170) for achieving an appropriate switching control of thethreshold voltage Vth.

<Reference Voltage Generator, Comparison Section>

FIG. 9 is a circuit diagram illustrating an example of a configurationof the reference voltage generator 150 and the comparison section 160.The reference voltage generator 150 includes a current source 151 and aresistor 152 (resistance: R152). The comparison section 160 includes acomparator 161.

The current source 151 is connected between the application terminal ofthe boosted voltage VG and the resistor 152, and outputs the secondcurrent Iset (more precisely, a variable current equivalent to thesecond current Iset) which is generated by the second current generator120.

The resistor 152 is connected between the current source 151 and theapplication terminal of the output voltage Vo (=the external terminalT2), and is a current/voltage conversion element that generates thereference voltage VIset (=Iset×R152) in accordance with the secondcurrent Iset.

A power supply terminal of the comparator 161 is connected to theapplication terminal of the boosted voltage VG. A reference potentialterminal of the comparator 161 is connected to the application terminalof the output voltage Vo (the external terminal T2). A non-invertinginput terminal (+) of the comparator 161 is connected to the applicationterminal of the sense voltage Vs. An inverting input terminal (−) of thecomparator 161 is connected to an application terminal of the referencevoltage VIset. The comparator 161 connected in this fashion compares thesense voltage Vs with the reference voltage VIset, and thereby generatesa comparison signal VCMP. The comparison signal VCMP becomes low levelwhen the sense voltage Vs is lower than the reference voltage VIset, andbecomes high level when the sense voltage Vs is higher than thereference voltage VIset.

Here, the resistance R152 of the resistor 152 is switched to one of afirst resistance Rdet1 and a second resistance Rdet2 (Rdet1>Rdet2) inaccordance with the comparison signal VCMP. More specifically, theresistance R152 of the resistor 152 is the first resistance Rdet1 whenthe comparison signal VCMP is at low level, and is the second resistanceRdet2 when the comparison signal control signal is at high level. Bythus switching and controlling the resistance R152, it is possible togive a hysteresis characteristic to the comparison section 160.

<Threshold Controller>

FIG. 10 is a circuit diagram illustrating an example of a configurationof the threshold controller 170. The threshold controller 170 includes acomparator 171, a current source 172, a level shifter 173, an RSflip-flop 174, a discharge controller 175, an NMOSFET 176, a capacitor177, and an external terminal DLY.

A power supply terminal of the comparator 171 is connected to theapplication terminal of the internal power supply voltage Vreg. Areference potential terminal of the comparator 171 is connected to theground terminal GND. A non-inverting input terminal (+) of thecomparator 171 is connected to the external terminal DLY (an applicationterminal of a charge voltage Vd). An inverting input terminal (−) of thecomparator 171 is connected to an application terminal of a mask periodexpiration voltage Vdref. The comparator 171 connected in this fashioncompares the charge voltage Vd with the mask period expiration voltageVdref, and thereby generates an internal signal Sx. The internal signalSx becomes high level when the charge voltage Vd is higher than the maskperiod expiration voltage Vdref, and becomes low level when the chargevoltage Vd is lower than the mask period expiration voltage Vdref.

The current source 172 is connected between the application terminal ofthe internal power supply voltage Vreg and the external terminal DLY,and generates a predetermined charge current Id. Here, whether or notthe current source 172 is operable is controlled in accordance with aninternal signal Sy (=corresponding to the comparison signal VCMP afterbeing subjected to level shifting). More specifically, the currentsource 172 is in an operating state when the internal signal Sy is athigh level, and is in a non-operating state when the internal signal Syis at low level.

The level shifter 173 level-shifts the comparison signal VCMP which ispulse-driven between the boosted voltage VG and the output voltage Vo tothereby generates the internal signal Sy which is pulse-driven betweenthe internal power supply voltage Vreg and a ground voltage GND.Accordingly, when the comparison signal VCMP is at high level (=VG), theinternal signal Sy is also at high level (=Vreg), and when thecomparison signal VCMP is at low level (=Vo), the internal signal Sy isalso at low level (=GND).

The RS flip-flop 174 outputs the threshold control signal S170 from itsoutput terminal (Q) in accordance with the internal signal Sx inputtedto its set terminal (S) and the internal signal Sy inputted to its thereset terminal (R). More specifically, the RS flip-flop 174 sets thethreshold control signal S170 to high level at rising timing of theinternal signal Sx, while it resets the threshold control signal S170 tolow level at falling timing of the internal signal Sy.

The discharge controller 175 generates an internal signal Sz inaccordance with the internal signal Sx. More specifically, the dischargecontroller 175 causes the internal signal Sz to rise to high level atrising timing of the internal signal Sx, and keeps the internal signalSz at high level over a predetermined discharge period Tdchg.

The NMOSFET 176 is a discharge switch element which achieves aconducting/cutoff state between the external terminal DLY and the groundterminal GND (=between the two terminals of the capacitor 177) inaccordance with the internal signal Sz. Here, the NMOSFET 176 is turnedon when the internal signal Sz is at high level, and is turned off whenthe internal signal Sz is at low level.

The capacitor 177 is connected between the external terminal DLY and theground terminal GND, outside the semiconductor integrated circuit device1. When the NMOSFET 176 is in an off-state, if the charge current Id issupplied from the current source 172, the charge voltage Vd of thecapacitor 177 rises. On the other hand, when the NMOSFET 176 is in anon-state, the capacitor 177 is discharged via the NMOSFET 176, and thusthe charge voltage Vd lowers.

<Overcurrent Protection Circuit>

FIG. 11 is a timing chart illustrating an example of an overcurrentprotection operation, in which the external control signal Si, the firstcurrent Iref, the second current Iset, the sense voltage Vs, thecomparison signal VCMP, the charge voltage Vd, the internal signals Sxto Sz, the threshold control signal S170, the threshold voltage Vth, andthe status notification signal So are depicted in order from the top.

At time t11, when the external control signal Si rises to high level, anoperation of generating the first current Iref is started without delay.However, at time t11, since shutdown of the semiconductor integratedcircuit device 1 has not been cancelled and the NMOSFET 10 is maintainedin an off state, no output current Io flows in the NMOSFET 10.Accordingly, the sense voltage Vs is maintained at 0V.

At time t12, when a predetermined activation delay period Tdly (e.g., 5μs) elapses from time t11, the shutdown of the semiconductor integratedcircuit device 1 is cancelled. As a result, the NMOSFET 10 is turned onand the output current Io starts to flow, and thus the sense voltage Vsstarts to rise. Further, at time t12, operations of generating thesecond current Iset and the reference voltage VIset (in the figure,VIset=VthL) in accordance with the second current Iset are also started.Note that, at time t12, the sense voltage Vs is lower than the referencevoltage VIset, and thus the comparison signal VCMP becomes low level.Accordingly, the threshold control signal S170 becomes low level, andthus a state is reached where the internal set value VthH is selected asthe threshold voltage Vth.

At time t13, when the sense voltage Vs becomes higher than the referencevoltage VIset, the comparison signal VCMP becomes high level. As aresult, the internal signal Sy becomes high level, and thus the chargevoltage Vd starts to rise. Note that, at time t13, the charge voltage Vdis lower than the mask period expiration voltage Vdref, and thus theinternal signal Sx remains at low level. Accordingly, the thresholdcontrol signal S170 is maintained at low level, and thus the internalset value VthH remains selected as the threshold voltage Vth. Hence,even though the sense voltage Vs is higher than the external set valueVthL (=VIset), overcurrent protection is not applied.

At time t14, when the charge voltage Vd becomes higher than the maskperiod expiration voltage Vdref, the internal signal Sx becomes highlevel. Accordingly, the threshold control signal S170 is set to highlevel, and thus the threshold voltage Vth is switched to the externalset value VthL. As a result, from time t14, overcurrent protection isapplied so that the sense voltage Vs will not become higher than theexternal set value VthL. Further, when the internal signal Sx rises tohigh level, the internal signal Sz also becomes high level and remainsat high level over the predetermined discharge period Tdchg, and thusthe charge voltage Vd is discharged to 0 V. Here, the discharge periodTdchg is desirably a time (for example, 3 μs) that is shorter than theabove-mentioned activation delay period Tdly.

Thus, when the threshold voltage Vth is set to the internal set valueVthH, the threshold voltage Vth is switched to the external set valueVthL at a time point when a predetermined mask period Tmask (=time t13to time t14) elapses with the sense voltage Vs remaining higher than thereference voltage VIset. Accordingly, it is possible to achieveovercurrent protection suitable for the load 3.

On the other hand, although not illustrated in the figure, even when thesense voltage Vs is higher than the reference voltage VIset for aninstant, if the sense voltage Vs becomes lower than the referencevoltage VIset again before expiration of the mask period Tmask, thethreshold voltage Vth remains maintained at the internal set value VthH.Thus, no unintended overcurrent protection is applied, so that it ispossible to secure an instantaneous current on startup.

Furthermore, naturally, when the threshold voltage Vth is set to theinternal set value VthH, if the sense voltage Vs becomes higher than theinternal set value VthH, at that time point, without delay, overcurrentprotection is applied. Thus, it is possible to forcibly turn off theNMOSFET 10 as soon as abnormality such as a short circuit occurs in theload 3, and this helps prevent destruction of the semiconductorintegrated circuit device 1 itself.

Here, the mask period Tmask described above is a variable value that isadjustable as necessary by using the capacitor 177, which is connectedexternally. More specifically, the larger the capacitance of thecapacitor 177 is, the longer the mask period Tmask is, and the smallerthe capacitance of the capacitor 177 is, the shorter the mask periodTmask is. However, as the mask period Tmask becomes longer, overcurrentprotection using the external set value VthL is started with more delay.Thus, it is desirable to set the mask period Tmask to a minimumnecessary length, considering duration of the instantaneous current atthe start-up.

Furthermore, it is also possible to appropriately select whether or notto provide the mask period Tmask depending on the purpose for which thesemiconductor integrated circuit device 1 is used (the kind of the load3). For example, when the external terminal DLY is open, the mask periodTmask is substantially zero, which is equivalent to a case where onlythe external set value VthL is provided. Furthermore, for example, whenthe external terminal DLY is short-circuited with the ground terminalGND, the mask period Tmask is infinite, which is equivalent to a casewhere only the internal set value VthH is provided.

At time t15, when the sense voltage Vs becomes lower than the referencevoltage VIset, the comparison signal VCMP becomes low level, an thus theinternal signal Sy becomes low level. As a result, the threshold controlsignal S170 is reset to low level, and thus the threshold voltage Vth isswitched to the internal set value VthH.

Thus, when the threshold voltage Vth is set to the external set valueVthL, at a time point when the sense voltage Vs becomes lower than thereference voltage VIset, the threshold voltage Vth is switched to theinternal set value VthH. That is, when the overcurrent protectionoperation using the external set value VthL is canceled, the overcurrentprotection circuit 71 is returned to its initial state at startup.

At time t16, when the external control signal Si falls to low level, thesemiconductor integrated circuit device 1 is shut down, and thus theabove series of operation is ended.

Here, with attention on the status notification signal So, during aperiod when no overcurrent is detected (a period excluding from time t14to time t15), an output detection voltage V80 (see also the broken linein the figure), which corresponds to a detection result of the outputcurrent Io, is selectively outputted. On the other hand, during a periodwhen an overcurrent is detected (a period from time t14 to time t15),instead of the output detection voltage V80, a constant voltage V90,which corresponds to the abnormality flag, is selectively outputted.

FIG. 12 is a flowchart illustrating an example of a threshold switchingoperation. When the flow starts, first, in step S101, the thresholdvoltage Vth is set to the internal set value VthH(=Iref×Rref1)(corresponding to time t12 of FIG. 11).

Next, in step S102, a determination is made on whether or not the sensevoltage Vs is higher than the reference voltage VIset. When aYes-determination is made here, the flow proceeds to step S103. On theother hand, when a No-determination is made, the flow returns to stepS102, and the determination in this step is repeated (which correspondsto from time t12 to time t13 of FIG. 11).

In step S103, in response to the Yes-determination made in step S102,the capacitor 177 starts to be charged (which corresponds to time t13 ofFIG. 11).

Next, in step S104, a determination is made on whether or not the chargevoltage Vd is higher than the mask period expiration voltage Vdref. Whena Yes-determination is made here, the flow proceeds to step S105. On theother hand, when a No-determination is made, the flow returns to stepS104, and the determination in this step is repeated (which correspondsto from time t13 to time t14 of FIG. 11).

In step S105, in response to the Yes-determination made in step S104,the capacitor 177 is discharged. In step S106, the threshold voltage Vthis switched to the external set value VthL (=Iset×Rref2). Steps S105 andS106 correspond to time t14 of FIG. 11.

Next, in step S107, a determination is made on whether or not the sensevoltage Vs is lower than the reference voltage VIset. When aYes-determination is made here, the flow returns to step S101, and thethreshold voltage Vth is switched to the internal set value VthH(=Iref×Rref1) again (corresponding to time t15 of FIG. 11). On the otherhand, when a No-determination is made, the flow returns to step S107,and the determination in this step is repeated (corresponding to fromtime t14 to time t15 of FIG. 11).

Usage Example

FIG. 13 is a schematic diagram illustrating a first usage example of theovercurrent protection circuit 71. For example, when the load 3 is abulb lamp, as indicated by the solid line in the figure, the outputcurrent Io flowing as an instantaneous current on startup is larger thanthat in a steady-state operation. However, with the above-described maskperiod Tmask set appropriately, this instantaneous current can beexcluded from detection targets, so that no overcurrent protection isunintendedly applied. That is, on startup, when an excessiveinstantaneous current flows, the output current Io and a first set valueIocpH are compared with each other, whereas the output current Io and asecond set value IocpL are compared with each other in the steady-stateoperation. Accordingly, the output current Io has a driving zone that isindicated in the figure with hatching.

FIG. 14 is a schematic diagram illustrating a second usage example ofthe overcurrent protection circuit 71. For example, when the load 3 is amotor, as indicated by the solid line in the figure, the output currentIo flowing as an instantaneous current when the motor is locked islarger than that in a steady-state operation. However, with theabove-described mask period Tmask set appropriately, this instantaneouscurrent can be excluded from detection targets, and thus no overcurrentprotection is unintendedly applied. That is, when the motor is lockedand an excessive instantaneous current flows, the output current Io andthe first set value IocpH are compared with each other, whereas theoutput current Io and the second set value IocpL are compared with eachother in the steady-state operation. Accordingly, the output current Iohas a driving zone that is indicated in the figure with hatching.

Effects and Advantages

As has been discussed above, in the overcurrent protection circuit 71,as the overcurrent set value Iocp to be compared with the output currentIo, there are prepared two values, which are, namely, the first setvalue IocpH and the second set value IocpL, and furthermore, thepredetermined mask period Tmask is provided as a moratorium periodbefore switching from the first set value IocpH to the second set valueIocpL.

By adopting this configuration, it is possible to achieve both thesecuring of an instantaneous current and overcurrent protection suitablefor the load 3. In particular, during the steady-state operation of theload 3, the second set value IocpL, which is sufficiently lower than thefirst set value IocpH, is compared with the output current Io, and thishelps prevent a large current that is too much larger than a drivecurrent of the load 3 from continuously flowing as the output currentIo. This allows a harness to be connected to the load 3 to have adiameter that is smaller than diameters of conventional harnesses.

Furthermore, with the overcurrent protection circuit 71, there is noneed of performing overcurrent protection suitable for the load 3 in theECU 2, and this makes it possible to reduce burden on the ECU 2(=constant monitoring of the output current Io, for example), and thusto achieve the ECU 2 without a microcomputer.

Semiconductor Integrated Circuit Device (Second Embodiment)

FIG. 15 is a block diagram illustrating a second embodiment of thesemiconductor integrated circuit device 1. The semiconductor integratedcircuit device 1 of the present embodiment is based on the firstembodiment (FIG. 1), but, for the purpose of separately drivingtwo-channel loads 3X and 3Y, it has the above-described components (thefunctional blocks 10 to 90, the external terminals T1 to T4, variousvoltages, currents and signals, etc.) for each channel.

The letter “X” is appended to ends of symbols for components related tothe driving of the load 3X, while the letter “Y” is appended to ends ofsymbols for components related to the driving of the load 3Y, but theiroperations and functions are basically the same as those of theabove-mentioned components denoted by symbols without the letter “X” or“Y” at their ends. For example, operations and functions of the NMOSFETs10X and 10Y are basically the same as those of the NMOSFET 10 describedabove. This applies also to the other components. Hence, unless there issomething especially noteworthy, overlapping descriptions of theoperations and functions of the components will be omitted. Furthermore,the output current detector 80 and the signal output section 90 are notclearly indicated in the figure, and these function blocks will bedescribed later.

In the semiconductor integrated circuit device 1 of the presentembodiment, which is capable of separately driving the two-channel loads3X and 3Y, there is a case where startup takes place at differenttimings for the different channels. Thus, in order to achieve both thesecuring of an instantaneous current and overcurrent protection suitablefor the load of each channel, the mask period Tmask described aboveneeds to be set correctly independent of the difference in startuptiming.

The simplest configuration to achieve this is obtained by preparing theabove-described overcurrent protection circuit 71 (see FIG. 4) for eachof the two channels, that is, by providing overcurrent protectioncircuits 71X and 71Y in parallel with each other. However, such aconfiguration requires two external terminals DLY for setting the maskperiod Tmask, which may necessitate the changing of the package of thesemiconductor integrated circuit device 1 or invite an increase in costof the semiconductor integrated circuit device 1.

To prevent such disadvantages, an overcurrent protection circuit 71 isproposed below which is capable of correctly setting the mask periodTmask for each channel without adding another external terminal DLY.

FIG. 16 is a block diagram illustrating an example of a configuration ofthe overcurrent protection circuit 71 having a two-channelconfiguration. The overcurrent protection circuit 71 of thisconfiguration example includes a first current generator 110, a secondcurrent generator 120, threshold voltage generators 130X and 130Y,overcurrent detectors 140X and 140Y, reference voltage generators 150Xand 150Y, comparison sections 160X and 160Y, and a threshold controller170.

Of the above components, the first current generator 110, the secondcurrent generator 120, the threshold voltage generator 130X, theovercurrent detector 140X, the reference voltage generator 150X, thecomparison section 160X, and the threshold controller 170 function as anovercurrent protection circuit 71X for a first channel.

On the other hand, of the above components, the first current generator110, the second current generator 120, the threshold voltage generator130Y, the overcurrent detector 140Y, the reference voltage generator150Y, the comparison section 160Y, and the threshold controller 170function as an overcurrent protection circuit 71Y for a second channel.

In this fashion, in the overcurrent protection circuit 71 of the presentconfiguration example, the first current generator 110, the secondcurrent generator 120, and the threshold controller 170 are shared bythe first channel and the second channel.

The first current generator 110 generates the first current Iref andoutputs it to the threshold voltage generators 130X and 130Y. The firstcurrent Iref has a current value that is fixed inside the semiconductorintegrated circuit device. The first current generator 110 is configuredbasically as illustrated in FIG. 5 referred to above. As means foroutputting the first current Iref to both of the threshold voltagegenerators 130X and 130Y, there may be used, for example, a currentmirror having current output terminals for two systems.

The second current generator 120 generates the second current Iset andoutputs it to the threshold voltage generators 130X and 130Y. The secondcurrent Iset has a current value that is adjustable as necessary fromoutside the semiconductor integrated circuit device 1. The secondcurrent generator 120 is configured basically as illustrated in FIG. 6referred to above. As means for outputting the second current Iset toboth of the threshold voltage generators 130X and 130Y, there may beused, for example, a current mirror having current output terminals fortwo systems.

The threshold voltage generator 130X switches, in accordance with athreshold control signal S170X, whether to set a threshold voltage VthXto an internal set value VthXH or to an external set value VthXL (whereVthXH>VthXL). The internal set value VthXH is a fixed value(=corresponding to the first set value) that is set in accordance withthe first current Iref. On the other hand, the external set value VthXLis a variable value (=corresponding to the second set value) that is setin accordance with the second current Iset.

The threshold voltage generator 130Y switches, in accordance with athreshold control signal S170Y, whether to set a threshold voltage VthYto an internal set value VthYH or to an external set value VthYL (whereVthXH>VthXL). The internal set value VthYH is a fixed value(=corresponding to the third set value) that is set in accordance withthe first current Iref. On the other hand, the external set value VthYLis a variable value (=corresponding to the fourth set value) that is setin accordance with the second current Iset.

The overcurrent detector 140X compares a sense voltage VsX, which is inaccordance with an output current IoX, with the threshold voltage VthX,and thereby generates an overcurrent protection signal S71X.

The overcurrent detector 140Y compares a sense voltage VsY, which is inaccordance with an output current IoY, with the threshold voltage VthY,and thereby generates an overcurrent protection signal S71Y.

The reference voltage generator 150X generates a reference voltageVIsetX (=corresponding to a first reference value) in accordance withthe second current Iset.

The reference voltage generator 150Y generates a reference voltageVIsetY (=corresponding to a second reference value) in accordance withthe second current Iset.

The comparison section 160X compares the sense voltage VsX with thereference voltage VIsetX, and thereby generates a comparison signalVCMPX.

The comparison section 160Y compares the sense voltage VsY with thereference voltage VIsetY, and thereby generates a comparison signalVCMPY.

The threshold controller 170 monitors both the comparison signals VCMPXand VCMPY, and thereby generates the threshold control signals S170X andS170Y.

Here, the threshold control signal S170X becomes low level when theinternal set value VthXH should be selected as the threshold voltageVthX, and becomes high level when the external set value VthXL should beselected as the threshold voltage VthX.

On the other hand, the threshold control signal S170Y becomes low levelwhen the internal set value VthYH should be selected as the thresholdvoltage VthY, and becomes high level when the external set value VthYLshould be selected as the threshold voltage VthY.

Threshold Controller (First Example)

FIG. 17 is a block diagram illustrating a first example of the thresholdcontroller 170. The threshold controller 170 of the present example isbased on FIG. 10, which has been referred to above, and, as means forachieving a two-channel configuration, the threshold controller 170includes the comparator 171, the current source 172, level shifters 173Xand 173Y, RS flip-flops 174X and 174Y, the discharge controller 175, theNMOSFET 176, the capacitor 177, a charge controller 178, and theexternal terminal DLY.

The comparator 171 compares the charge voltage Vd (=a charge voltage ofthe capacitor 177 which appears at the external terminal DLY) inputtedto its non-inverting input terminal (+) with the mask period expirationvoltage Vdref inputted to its inverting input terminal (−), and therebygenerates the internal signal Sx. The internal signal Sx becomes highlevel when the charge voltage Vd is higher than the mask periodexpiration voltage Vdref, and becomes low level when the charge voltageVd is lower than the mask period expiration voltage Vdref. This issimilar to what is described in FIG. 10, which has been referred toabove.

The current souce 172 generates the charge current Id in accordance witha charge control signal S178. More specifically, the current source 172outputs the charge current Id when the charge control signal S178 is athigh level, and stops the charge current Id when the charge controlsignal S178 is at low level.

The level shifter 173X level-shifts the comparison signal VCMPX, andthereby generates an internal signal SyX.

The level shifter 173Y level-shifts the comparison signal VCMPY, andthereby generates an internal signal SyY.

The RS flip-flop 174X outputs the threshold control signal S170X fromits output terminal (Q) in accordance with the internal signal Sxinputted to its set terminal (S) and the internal signal SyX inputted toits the reset terminal (R). More specifically, the RS flip-flop 174Xsets the threshold control signal S170X to high level at rising timingof the internal signal Sx, while it resets the threshold control signalS170X to low level at falling timing of the internal signal SyX.

The RS flip-flop 174Y outputs the threshold control signal S170Y fromits output terminal (Q) in accordance with the internal signal Sxinputted to its set terminal (S) and the internal signal SyY inputted toits reset terminal (R). More specifically, the RS flip-flop 174Y setsthe threshold control signal S170Y to high level at rising timing of theinternal signal Sx, while it resets the threshold control signal S170Yto low level at falling timing of the internal signal SyY.

The discharge controller 175 generates the internal signal Sz inaccordance with the internal signal Sx. More specifically, the dischargecontroller 175 causes the internal signal Sz to rise to high level atrising timing of the internal signal Sx, and keeps the internal signalSz at high level over the predetermined discharge period Tdchg. This issimilar to what is described in FIG. 10, which has been referred toabove.

The NMOSFET 176 is a discharge switch element which achieves aconducting/cutoff state between the external terminal DLY and the groundterminal GND (=between the two terminals of the capacitor 177) inaccordance with the internal signal Sz. Here, the NMOSFET 176 is turnedon when the internal signal Sz is at high level, and is turned off whenthe internal signal Sz is at low level. This also is similar to what isdescribed in FIG. 10, which has been referred to above.

The capacitor 177 is connected between the external terminal DLY and theground terminal GND, outside the semiconductor integrated circuit device1. When the NMOSFET 176 is OFF, if the charge current Id is suppliedfrom the current source 172, the charge voltage Vd of the capacitor 177rises. On the other hand, when the NMOSFET 176 is ON, the capacitor 177is discharged via the NMOSFET 176, and thus the charge voltage Vd falls.This also is similar to what is described in FIG. 10, which has beenreferred to above.

The charge controller 178 generates the charge control signal S178 inaccordance with both the internal signals SyX and SyY (thus thecomparison signals VCMPX and VCMPY). The charge control signal S178basically rises to high level (=a logic level at a time of charging) atrising timing of the internal signal SyX or SyY.

FIG. 18 is a timing chart illustrating the threshold switching operationof the first example, in which the sense voltages VsX and VsY, thecomparison signals VCMPX and VCMPY (equivalent to the internal signalsSyX and SyY), the charge voltage Vd, the internal signals Sx and Sz, thethreshold control signals S170X and S170Y, and the threshold voltagesVthX and VthY are depicted in order from the top.

At time t21, when the NMOSFET 10X is turned on, the sense voltage VsXstarts to rise. However, at time t21, the sense voltage VsX is lowerthan the reference voltage VIsetX, and thus the comparison signal VCMPX(=the internal signal SyX) becomes low level. Accordingly, the thresholdcontrol signal S170X becomes low level, and thus a state is reachedwhere the internal set value VthXH is selected as the threshold voltageVthX. Here, at time t21, the NMOSFET 10Y remains OFF, and the sensevoltage VsY is maintained at 0V.

At time t22, when the sense voltage VsX becomes higher than thereference voltage VIsetX, the comparison signal VCMPX (=the internalsignal SyX) becomes high level, and the charge voltage Vd starts torise. However, since the charge voltage Vd is lower than the mask periodexpiration voltage Vdref at time t22, the internal signal Sx remains atlow level. Accordingly, the threshold control signal S170X is maintainedat low level, and the internal set value VthH remains selected as thethreshold voltage VthX. Thus, even though the sense voltage VsX ishigher than the external set value VthXL (=VIsetX), no overcurrentprotection is applied. Here, at time t22, the NMOSFET 10Y remains OFF,and the sense voltage VsY is maintained at 0V.

At time t23, the NMOSFET 10Y is turned on, and the sense voltage VsYstarts to rise. Here, at time t23, the sense voltage VsY is lower thanthe reference voltage VIsetY, and thus the comparison signal VCMPY (=theinternal signal SyY) becomes low level. Accordingly, the thresholdcontrol signal S170Y becomes low level, and thus a state is reachedwhere the internal set value VthYH is selected as the threshold voltageVthY.

At time t24, when the charge voltage Vd becomes higher than the maskperiod expiration voltage Vdref, the internal signal Sx becomes highlevel. At time t24, the comparison signal VCMPX (=the internal signalSyX) has already become high level (=a logic level at a time of resetcancellation). Accordingly, the threshold control signal S170X is set tohigh level, and the threshold voltage VthX is switched to the externalset value VthXL. As a result, at time t24, overcurrent protection startsto be applied so that the sense voltage VsX will not become higher thanthe external set value VthXL. Further, when the internal signal Sx risesto high level, the internal signal Sz also becomes high level andremains at high level over the predetermined discharge period Tdchg, andthus the charge voltage Vd is discharged to 0 V.

That is, with attention on the threshold voltage VthX, when thethreshold voltage VthX is set to the internal set value VthXH, thethreshold voltage VthX is switched to the external set value VthXL at atime point when a predetermined mask period Tmask (=time t22 to timet24) elapses with the sense voltage VsX remaining higher than thereference voltage VIsetX. Accordingly, it is possible to achieveovercurrent protection suitable for the load 3X.

On the other hand, at time t24, the comparison signal VCMPY (=theinternal signal SyY) is maintained at high level (=a logic level at atime of reset). Accordingly, even when the internal signal Sx rises tohigh level, the threshold control signal S170Y is maintained at lowlevel, and thus the internal set value VthYH remains selected as thethreshold voltage VthY.

At time t25, when the sense voltage VsY becomes higher than thereference voltage VIsetY, the comparison signal VCMPY (=the internalsignal SyY) becomes high level, and thus the charge voltage Vd starts torise again. However, since the charge voltage Vd is lower than the maskperiod expiration voltage Vdref at time t25, the internal signal Sxremains at low level. Accordingly, the threshold control signal S170Y ismaintained at low level, and the internal set value VthYH remainsselected as the threshold voltage VthY. Thus, even though the sensevoltage VsY is higher than the external set value VthYL (=VIsetY), noovercurrent protection is applied.

In the following description, a period between rising timing of thecomparison signal VCMPX and rising timing of the comparison signal VCMPY(=a period between first-channel startup timing and second-channelstartup timing) will be referred to as a shift period Tshift.

At time t26, when the sense voltage VsX becomes lower than the referencevoltage VIsetX, the comparison signal VCMPX (=the internal signal SyX)becomes low level. As a result, the threshold control signal S170X isreset to low level, and thus the threshold voltage VthX is switched tothe internal set value VthXH.

That is, with attention on the threshold voltage VthX, when thethreshold voltage VthX is set to the external set value VthXL, at a timepoint when the sense voltage VsX becomes lower than the referencevoltage VIsetX, the threshold voltage VthX is switched to the internalset value VthXH.

At time t27, when the charge voltage Vd becomes higher than the maskperiod expiration voltage Vdref, the internal signal Sx becomes highlevel. Further, at time t27, the comparison signal VCMPY (=the internalsignal SyY) has already become high level (=a logic level at a time ofreset cancellation). Accordingly, the threshold control signal S170Y isset to high level, and the threshold voltage VthY is switched to theexternal set value VthXL. As a result, from time t27, overcurrentprotection is applied so that the sensing voltage VsY will not becomehigher than the external set value VthYL. Also, when the internal signalSx becomes high level, the internal signal Sz also becomes high leveland remains at high level over the predetermined discharge period Tdchg,and thus the charge voltage Vd is discharged to 0 V.

That is, with attention on the threshold voltage VthY, when thethreshold voltage VthY is set to the internal set value VthYH, thethreshold voltage VthY is switched to the external set value VthYL at atime point when a predetermined mask period Tmask (=time t25 to timet27) elapses with the sense voltage VsY remaining higher than thereference voltage VIsetY. Accordingly, it is possible to achieveovercurrent protection suitable for the load 3Y.

Here, at time t27, the comparison signal VCMPX(=the internal signal SyX)has already fallen to low level (=the logic level at the time of reset).Accordingly, even when the internal signal Sx rises to high level, thethreshold control signal S170X is maintained at low level, and thus theinternal set value VthXH remains selected as the threshold voltage VthX.

At time t28, when the sense voltage VsY becomes lower than the referencevoltage VIsetY, the comparison signal VCMPY(=the internal signal SyY)becomes low level. As a result, the threshold control signal S170Y isreset to low level, and thus the threshold voltage VthY is switched tothe internal set value VthYH.

That is, with attention on the threshold voltage VthY, when thethreshold voltage VthY is set to the external set value VthYL, at a timepoint when the sense voltage VsY becomes lower than the referencevoltage VIsetY, the threshold voltage VthY is switched to the internalset value VthYH.

As is clear from the above-described series of threshold switchingoperation, with the threshold controller 170 of the present example,without the need of adding another external terminal DLY, it is possibleto correctly set the mask period Tmask (from time t22 to time t23, andfrom time t25 to time t27) for each channel.

The above description has been given with reference to the presentfigure by taking as an example a case where Tshift>Tmask; in a casewhere Tshift≤Tmask by contrast, problems may occur in theabove-described series of threshold switching operation. Hereinafter, adetailed description will be given of such problems.

FIG. 19 is a timing chart illustrating problems that the first examplesuffers, in which there are depicted behaviors of the comparison signalsVCMPX and VCMPY, the internal signal Sx, and the threshold controlsignals S170X and S170YT, in order from the top, observed in a casewhere Tshift<Tmask.

In the example illustrated in the figure, where Tshift<Tmask, after thecomparison signal VCMPX rises to high level at time t31, the comparisonsignal VCMOY rises to high level at time t32 before the mask periodTmask elapses.

Accordingly, when the mask period Tmask elapses from time t31 and theinternal signal Sx rises to high level at time t33, not only thecomparison signal VCMPX but also the comparison signal VCMPY has alreadybecome high level. Thus, at time t33, the threshold control signalsS170X and S170Y simultaneously become high level.

In this case, a preceding channel, which is started up first, does nothave a problem in particular, but with respect to a subsequent channel,which is started up after the preceding channel, the mask period Tmaskis shorter by the length of the shift period Tshift, and this may makeit difficult to secure an instantaneous current. Hereinafter, a secondexample of the threshold controller 170 will be proposed which iscapable of solving this problem.

Threshold Controller (Second Example)

FIG. 20 is a block diagram illustrating a second example of thethreshold controller 170. The threshold controller 170 of the presentexample is based on the first example (FIG. 17) described above, and ischaracterized in that the discharge controller 175 accepts input of notonly the internal signal Sx but also the internal signals SyX and SyY(equivalent to the comparison signals VCMPX and VCMPY) and the thresholdcontrol signals S170X and S170Y. The following description will focus onthe configuration and the operation of the discharge controller 175.

FIG. 21 is a block diagram illustrating an example of a configuration ofthe discharge controller 175. The discharge controller 175 illustratedin the present figure includes an NOR operation unit NOR1, AND operationunits AND1 to AND3, an OR operation unit OR1, inverters INV1 to INV3, apulse generator PG1, a resistor R1, and a capacitor C1.

The NOR operation unit NOR1 performs a NOR operation on the thresholdcontrol signals S170X and S170Y, and thereby generates a logic signalSA. Accordingly, the logic signal SA becomes high level when thethreshold control signals S170X and S170Y are both at low level, andbecomes low level when at least one of the threshold control signalsS170X and S170Y is at high level.

The AND operation unit AND1 performs an AND operation on the internalsignals SyX and SyY, and thereby generates a logic signal SB.Accordingly, the logic signal SB becomes high level when the internalsignals SyX and SyY are both high level, and becomes low level when atleast one of the internal signals SyX and SyY is at low level.

The AND operation unit AND2 performs an AND operation on the logicsignals SA and SB, and thereby generates a logic signal SC. Accordingly,the logic signal SC becomes high level when the logic signals SA and SBare both at high level, and becomes low level when at least one of thelogic signals SA and SB is at low level.

The inverter INV1 inverts the logic of the logic signal SC, and therebygenerates an inverted logic signal SCB.

The resistor R1 and the capacitor C1 generate a logic signal SD havingan integrated waveform obtained by dulling the inverted logic signal SCBwith a predetermined time constant τ (=R×C).

The inverters INV2 and INV3 compare the logic signal SD with apredetermined threshold (=a logic inversion threshold of the invertersINV2 and INV3), and thereby generate a logic signal SE having arectangular waveform.

The AND operation unit AND3 performs an AND operation on the logicsignals SC and SE, and thereby generates a logic signal SF. Accordingly,the logic signal SF becomes high level when the logic signals SC and SEare both at high level, and becomes low level when at least one of thelogic signals SC and SE is at low level.

At rising timing of the internal signal Sx, the pulse generator PG1generates, in the logic signal SG, a one-shot pulse having apredetermined pulse width (=corresponding to the discharge periodTdchg).

The OR operation unit OR1 performs an OR operation on the logic signalsSF and SG, and thereby generates the internal signal Sz. Accordingly,the internal signal Sz becomes low level when the logic signals SF andSG are both at low level, and becomes high level when at least one ofthe logic signals SF and SG is at high level.

FIG. 22 is a timing chart illustrating the threshold switching operationof the second example, in which there are depicted behaviors of thecomparison signals VCMPX and VCMPY (equivalent to the internal signalsSyX and SyY), the logic signals SA to SG, the internal signal Sz, thecharge voltage Vd, the internal signal Sx, and the threshold controlsignals S170X and S170Y, in order from the top, observed in a case whereTshift<Tmask.

In the example illustrated in the present figure, after the comparisonsignal VCMPX rises to high level at time t41, the comparison signalVCMPY rises to high level at time t42 before the mask period Tmaskelapses. That is, at time t42, the charge voltage Vd has not reached themask period expiration voltage Vdref yet, and the internal signal Sx hasnot risen to high level yet.

Here, with attention on an internal operation of the dischargecontroller 175, at time t42, the threshold control signals S170X andS170Y are both at low level, and accordingly the logic signal SA is athigh level. At time t42, the comparison signals VCMPX and VCMPY (andthus the internal signals SyX and SyY) are both at high level, and thusthe logic signal SB rises to high level. Accordingly, the logic signalSC rises to high level, and the logic signal SD starts to fall with thetime constant τ. However, at time t42, the logic signal SD is higherthan the logic inversion threshold of the inverter INV2, and thus thelogic signal SE is maintained at high level.

Accordingly, at time t42, since the logic signals SC and SE both becomehigh level, the logic signal SF rises to high level, and thus theinternal signal Sz rises to high level. As a result, the charge voltageVd is discharged.

In this fashion, when, after one of the comparison signals VCMPX andVCMPY rises to high level and a charging operation of the capacitor 177is started, the other one of the comparison signals VCMPX and VCMPYrises to high level before the charge voltage Vd becomes higher than themask period expiration voltage Vdref, the capacitor 177 is discharged,and thus an operation of counting the mask period Tmask is reset.

Then, at time t43, when the logic signal SD becomes lower than the logicinversion threshold of the inverter INV2, the logic signal SE falls tolow level. As a result, the logic signal SF falls to low level, and thusthe internal signal Sz falls to low level, the discharge operationdescribed above is stopped and the charge voltage Vd starts to riseagain.

Here, a high level period of the logic signal SF (=from time t42 to timet43) corresponds to the discharge period Tdchg2 of the charge voltageVd. The discharge period Tdchg2 is settable arbitrarily in accordancewith the time constant τ of the resistor R1 and the capacitor C1, and itmay be set, for example, to be equal to the above-described dischargeperiod Tdchg (for example, 3 μs).

Then, at time t44, when the charge voltage Vd becomes higher than themask period expiration voltage Vdref, the internal signal Sx rises tohigh level. At this time point, not only the comparison signal VCMPX butalso the comparison signal VCMPY has already become high level. Thus, attime t44, the threshold control signal S170X and S170Y simultaneouslybecome high level.

Through the threshold switching operation described above, as to thethreshold control signal S170Y of the subsequent channel, the length ofthe mask period becomes equal to an original set length (=Tmask). On theother hand, as to the threshold control signal S170X of the precedingchannel, the length of the mask period becomes longer (=Tmask+a) thanthe original set length.

Here, at time t44, when the internal signal Sx rises to high level,since the one-shot pulse having the predetermined pulse width (=Tdchg)is generated in the logic signal SG, the internal signal Sz becomes highlevel, and the charge voltage Vd is discharged.

Further, at time t44, when the threshold control signals S170X and 5170Yeach rise to high level, the logic signal SA falls to low level, and thelogic signal SC falls to low level. As a result, the logic signal SDstarts to rise with the time constant τ, and at a time point when thelogic signal SD becomes higher than the logic inversion threshold of theinverter INV2, the logic signal SE rises to high level. However, at thistime point, the logic signal SC has already become low level, and thusthe logic signal SF is maintained at low level.

As described above, with the threshold controller 170 of the presentexample, even in a case where Tshift<Tmask, the mask period of thesubsequent channel is not reduced, and thus, there is no risk that itwill become difficult to secure an instantaneous current.

Here, the above description has been given with reference to the presentfigure by taking as an example a case where Tshift<Tmask; under acritical condition where Tshift=Tmask (or, Tshift≈Tmas) by contrast,even by adopting the second example, there is a risk of an unexpectedmalfunction. This problem will be described below in detail.

FIG. 23 is a timing chart illustrating a problem that the second examplemay have, in which there are depicted behaviors of the comparisonsignals VCMPX and VCMPY (equivalent to the internal signals SyX andSyY), the charge voltage Vd, the internal signal Sx, and the thresholdcontrol signals S170X and S170Y, in order from the top, observed in acase where Tshift=Tmask.

In the example illustrated in the figure, since Tshift=Tmask, after thecomparison signal VCMPX rises to high level at time t51, the comparisonsignal VCMPY rises to high level at time t52, simultaneously with thelapse of the mask period Tmask.

Here, if the above-described discharge operation (see time t42 in FIG.22) fails to be in time, so that the charge voltage Vd becomes higherthan the mask period expiration voltage Vdref and the internal signal Sxrises to high level, the threshold control signals S170X and S170Ysimultaneously become high level. As a result, the mask period of thesubsequent channel becomes zero, and thus it becomes impossible tosecure an instantaneous current. Hereinafter, a third example of thethreshold controller 170 will be proposed, which is capable of solvingthe problem.

Threshold Controller (Third Example)

FIG. 24 is a block diagram illustrating a third example of the thresholdcontroller 170. The threshold controller 170 of the present example isbased on the second example (FIG. 20) described above, and ischaracterized by being provided with delay sections 179X and 179Y.Hence, the same components as those in the second example will be giventhe same reference symbols as those in FIG. 20 and thereby overlappingdescriptions thereof will be omitted, and the following description willfocus on the delay sections 179X and 179Y.

The delay section 179X gives a delay to the internal signal SyX(equivalent to the comparison signal VCMPX), and thereby generates adelay signal SyXd. Here, the delay section 179X gives a delay only torising timing of the delay signal SyXd, and does not give a delay tofalling timing of the delay signal SyXd. More specifically, the delaysignal SyXd rises to high level with a delay of delay time td (3 μs, forexample) after the internal signal SyX rising to high level, and fallsto low level simultaneously with the internal signal SyX falling to lowlevel.

The delay section 179Y gives a delay to the internal signal SyY(equivalent to the comparison signal VCMPY), and thereby generates adelay signal SyYd. Here, the delay section 179Y gives a delay only torising timing of the delay signal SyYd, and does not give a delay tofalling timing of the delay signal SyYd. More specifically, the delaysignal SyYd rises to high level with a delay of delay time td after theinternal signal SyY rising to high level, and falls to low levelsimultaneously with the internal signal SyY falling to low level.

As a result of the additional provision of the delay section 179X and179Y, the delay signals SyXd and SyYd, instead of the internal signalsSyX ad SyY, are inputted to the RS flip-flops 174X and 174Y,respectively.

FIG. 25 is a timing chart illustrating the threshold switching operationof the third example, in which there are depicted behaviors of thecomparison signal VCMPX (equivalent to the internal signal SyX), thedelay signal SyXd, the comparison signal VCMPY (equivalent to theinternal signal SyY), the delay signal SyYd, the internal signal Sz, thecharge voltage Vd, the internal signal Sx, and the threshold controlsignals S170X and S170Y, in order from the top, observed in a case whereTshift=Tmask.

In the example illustrated in the figure, since Tshift=Tmask, after thecomparison signal VCMPX (=SyX) rises to high level at time t61, thecomparison signal VCMPY (=SyY) rises to high level at time t62,simultaneously with the lapse of the mask period Tmask. On the otherhand, the delay signals SyXd and SyYd each have risen to high level at atime point when the predetermined delay time td has elapsed from timet61 and time t62, respectively.

Here, at time t62, when the charge voltage Vd becomes higher than themask period expiration voltage Vdref, the internal signal Sx becomeshigh level. At this time, the delay signal SyXd has already risen tohigh level (=the logic level at a time of reset cancellation).Accordingly, the threshold control signal S170X is set to high level attime t62.

On the other hand, at time t62, the delay signal SyYd is stillmaintained at low level (=the logic level at a time of reset).Accordingly, even when the internal signal Sx rises to high level, thethreshold control signal S170Y is maintained reset to low level.

Further, when the internal signal Sx rises to high level, since theinternal signal Sz becomes high level and remains at high level over thepredetermined discharge period Tdchg, the charge voltage Vd isdischarged to 0 V. Then, when the internal signal Sz falls to low levelat time t63, the discharge operation described above is stopped, and thecharge voltage Vd starts to rise again.

At time t64, when the charge voltage Vd becomes higher than the maskperiod expiration voltage Vdref, the internal signal Sx rises to highlevel again. At this time, the delay signal SyYd has already risen tohigh level (=the logic level at a time of reset cancellation).Accordingly, the threshold control signal S170Y is set to high level attime t64.

Further, when the internal signal Sx rises to high level, since theinternal signal Sz becomes high level and remains at high level over thepredetermined discharge period Tdchg, the charge voltage Vd isdischarged to 0 V. Then, when the internal signal Sz falls to low levelat time t65, the discharge operation described above is stopped. Here,at this time point, since the charge operation has been completed forthe two channels, the charge voltage Vd does not start to rise again.

Then, at time t66, when the comparison signal VCMPX (=internal signalSyX) falls to low level, the delay signal SyXd also falls to low levelwithout delay. As a result, the threshold control signal S170X is resetto low level.

Likewise, at time t67, when the comparison signal VCMPY (=the internalsignal SyY) falls to low level, the delay signal SyYd also falls to lowlevel without delay. As a result, the threshold control signal S170Y isreset to low level.

In this fashion, in the threshold controller 170 of the present example,the threshold control signals S170X and S170Y are generated by using theinternal signal Sx and the delay signals SyXd and SyYd. Thus, whenTshift≤Tmask, the charge voltage Vd never fails to be discharged atrising timing of the comparison signals VCMPX and VCMPY, before thedelay signals SyXd and SyYd each rise to high level.

Accordingly, even under a critical condition where Tshift=Tmask, thethreshold control signal S170X and S170Y never simultaneously becomehigh level, and thus it is possible to set a correct mask period Tmaskfor each channel.

<Flowchart>

FIG. 26 is a flowchart illustrating an example of a two-channelthreshold switching operation. When the flow starts, first, in stepS201, the threshold voltage Vth* for a channel that has been started upis set to the internal set value Vth*H (here, “*” is at least either “X”or “Y”, this applies also to the following description) (correspondingto time t21 and time t23 in FIG. 18).

Next, in step S202, a determination is made on whether or not one of thecomparison signals VCMPX and VCMPY is at high level (that is, whether ornot only one channel has been started up). When a Yes-determination ismade here, the flow proceeds to step S203 (corresponding to time t22 inFIG. 18). On the other hand, when a No-determination is made, the flowproceeds to step S208.

In step S203, in response to the Yes-determination made in step S202,the capacitor 177 starts to be charged (corresponding to time t22 inFIG. 18).

Next, in step S204, a determination is made on whether or not the chargevoltage Vd is higher than the mask period expiration voltage Vdref. Whena Yes-determination is made here, the flow proceeds to step S205(corresponding to time t24 in FIG. 18). On the other hand, when aNo-determination is made, the flow returns to step S204, and thedetermination in this step is repeated (corresponding to from time t22to time t24 in FIG. 18).

In step S205, in response to the Yes-determination made in step S204,the capacitor 177 is discharged. In step S206, the threshold voltageVth* for the channel that has been started up is switched to theexternal set value Vth*L. Steps S205 and S206 correspond to time t24 inFIG. 18.

Next, in step S207, a determination is made on whether or not the sensevoltage Vs* for the channel that has been started up is lower than thereference voltage VIset*. When a Yes-determination is made here, theflow returns to step S201, and the threshold voltage Vth* is switched tothe internal set value Vth*H again (corresponding to time t26 of FIG.18). On the other hand, when a No-determination is made, the flowreturns to step S207, and the determination in this step is repeated(corresponding to from time t24 to time t26 in FIG. 18).

On the other hand, in step S208, in response to the No-determinationmade in step S202, a determination is made on whether or not thecomparison signals VCMPX and VCMPY are both at high level (that is,whether or not the two channels have both been started up). When aYes-determination is made here, the flow proceeds to step S209(corresponding to time t23 in FIG. 18, time t42 in FIG. 22, or time t62in FIG. 25). On the other hand, when a No-determination is made, neitherof the channels has been started up, and thus the flow returns to stepS201.

In step S209, in response to the Yes-determination made in step S208, adetermination is made on whether or not one of the threshold signalsS170X and 5170Y is at high level (that is, whether or not the thresholdvoltage Vth* for the preceding channel has already been switched to theexternal set value Vth*L). When a Yes-determination is made here, theflow proceeds to step S203, and in steps S203 to S207, the thresholdswitching operation for the subsequent channel is performed(corresponding to from time t25 to time t28 in FIG. 18). On the otherhand, when a No-determination is made, the flow proceeds to step S210.

In step S210, in response to the No-determination made in step S209, adetermination is made on whether or not the threshold signals S170X andS170Y are both at low level (that is, whether or not the startup timingof the subsequent channel has come before a lapse of the mask periodTmask of the preceding channel. When a Yes-determination is made here,the flow proceeds to step S211 (corresponding to time t42 in FIG. 22).On the other hand, when a No-determination is made, the flow proceeds tostep S214.

In step S211, in response to the Yes-determination made in step S210,the capacitor 177 is discharged, and then starts to be charged again(corresponding to from time t 42 to time t43 in FIG. 22).

Next, in step S212, a determination is made on whether or not the chargevoltage Vd is higher than the mask period expiration voltage Vdref. Whena Yes-determination is made here, the flow proceeds to step S213(corresponding to time t44 in FIG. 22). On the other hand, when aNo-determination is made, the flow returns to step S212, and thedetermination in this step is repeated (corresponding to from time t43to time t44 in FIG. 22).

In step S213, in response to the Yes-determination made in step S212,the capacitor 177 is discharged. In step S214, the threshold voltagesVthX and VthYL for the two channels are switched to the external setvalues VthXL and VthYL. Steps S205 and S206 correspond to time t44 inFIG. 22.

Next, in step S215, a determination is made on whether or not the sensevoltages VsX and VsY for the two channels are lower than the referencevoltages VIsetX and VIsetY. When a Yes-determination is made here, theflow returns to step S201, into a state of waiting for a next startup.On the other hand, when a No-determination is made, the flow returns tostep S215, and the determination in this step is repeated.

<Multiplexer>

FIG. 27 is a block diagram illustrating an example in which amultiplexer is introduced as an output stage of the status notificationsignal So along with the two-channelization of the semiconductorintegrated circuit device 1, which has been described so far. In thesemiconductor integrated circuit device 1 of this configuration example,output current detectors 80X and 80Y, signal output sections 90X and90Y, a multiplexer 100, and an external terminal T5 are integrated.

The output current detector 80X generates a sense current IsX′ inaccordance with an output current IoX, and outputs the resulting sensecurrent IsX′ to the signal output section 90X.

The output current detector 80Y generates a sense current IsY′ inaccordance with an output current IoY, and outputs the resulting sensecurrent IsY′ to the signal output section 90Y.

The signal output section 90X includes a selector 91X which, based on anoutput selection signal S2X inputted from a control logic section 40X,selectively outputs one of the sense current IsX′ (=corresponding to aresult of detection of the output current IoX) and a fixted voltage V90(=corresponding to an abnormality flag) as a first status notificationsignal SoX. Here, the selector 91X selectively outputs the sense currentIsX′ as the first status notification signal SoX when the outputselection signal S2X is at a logic level (for example, low level) ofwhen no abnormality has been detected, and outputs the fixed voltage V90as the first status notification signal SoX when the output selectionsignal S2X is at a logic level (for example, high level) of when anabnormality has been detected.

The signal output section 90Y includes a selector 91Y which selectivelyoutputs one of the sense current IsY′ (=corresponding to a result ofdetection of the output current IoY) and a fixed voltage V90(=corresponding to an abnormality flag) as a second status notificationsignal SoY, based on an output selection signal S2Y inputted from acontrol logic section 40Y. Here, the selector 91Y selectively outputsthe sense current IsY′ as the second status notification signal SoY whenthe output selection signal S2Y is at a logic level (for example, lowlevel) of when no abnormality is detected, and outputs the fixed voltageV90 as the second status notification signal SoY when the outputselection signal S2Y is at a logic level (for example, high level) ofwhen an abnormality is detected.

In accordance with an output selection signal SEL inputted to theexternal terminal T5, the multiplexer 100 selectively outputs one of thefirst status notification signal SoX (=the sense current IsX′ or thefixed voltage V90) and a second status notification signal SoY (=thesense current IsY′ or the fixed voltage V90) to the external terminalT4.

In a case where the sense current IsX′ is selectively outputted to theexternal terminal T4, an output detection voltage V80X (=IsX′xR4)obtained by current-voltage conversion of the sense current IsX′ by theexternal sense resistor 4 is transmitted as the status notificationsignal So to the ECU 2. Here, the larger the output current IoX is, thehigher the output detection voltage V80X becomes, and the smaller theoutput current IoX is, the lower the output detection voltage V80Xbecomes.

In a case where the sense current IsY′ is selectively outputted to theexternal terminal T4, an output detection voltage V80Y (=IsY′×R4)obtained by current/voltage conversion of the sense current IsY′ by theexternal sense resistor 4 is transmitted as the status notificationsignal So to the ECU 2. Here, the larger the output current IoY is, thehigher the output detection voltage V80Y becomes, and the smaller theoutput current IoY is, the lower the output detection voltage V80Ybecomes.

On the other hand, in a case where the fixed voltage V90 is selectivelyoutputted to the external terminal T4, the fixted voltage V90 istransmitted as the status notification signal So to the ECU 2. Here, thefixed voltage V90 may be set to a voltage value higher than the upperlimit values of the output detection voltages V80X and V80Y.

With the introduction of the multiplexer 100 operating as describedabove, it is possible to externally monitor both the detection resultsof the output currents IoX and IoY and the abnormality flag for anarbitrary channel.

<Application to Vehicle>

FIG. 28 is an external view of a vehicle, illustrating an example of aconfiguration of the vehicle. A vehicle X of the present configurationexample has mounted therein a battery (not shown in the figure) andseveral electronic apparatuses X11 to X18 which operate with powersupplied from the battery. Here, for convenience of illustration,mounting positions of the electronic apparatuses X11 to X18 in thisfigure may be different from actual positions.

The electronic apparatus X11 is an engine control unit which performsengine-related control (injection control, electronic throttle control,idling control, oxygen sensor heater control, auto cruise control,etc.).

The electronic apparatus X12 is a lamp control unit which performsON/OFF control of an HID (high intensity discharged lamp) or a DRL(daytime running lamp).

The electronic apparatus X13 is a transmission control unit whichperforms transmission-related control.

The electronic apparatus X14 is a body control unit which performscontrol related to motion of the vehicle X (ABS (anti-lock brake system)control, EPS (electric power steering) control, electronic suspensioncontrol, etc.).

The electronic apparatus X15 is a security control unit that performscontrol of driving of a door lock, a security alarm, etc.

The electronic apparatus X16 is an electronic apparatus incorporated inthe vehicle X at the factory shipping stage as standard equipment or afactory-installed option, such as a wiper, an electric door mirror, apower window, a damper (a shock absorber), an electric sunroof, anelectric seat, etc.

The electronic apparatus X17 is an electronic apparatus that isoptionally mounted on the vehicle X as a user option such as anin-vehicle A/V (audio/visual) device, a car navigation system, an ETC(electronic toll collection system), etc.

The electronic apparatus X18 is an electronic apparatus, such as anin-vehicle blower, an oil pump, a water pump, a battery cooling fan,etc., which each include a high-withstanding-voltage motor.

Note that the semiconductor integrated circuit device 1, the ECU 2, andthe load 3 described above can be incorporated in any of the electronicapparatuses X11 to X18.

Other Modified Examples

In the above embodiments, the descriptions have been given by taking anin-vehicle high-side switch IC as an example, but application targets ofthe invention disclosed herein are not limited thereto; for example, thepresent invention disclosed herein is widely applicable not only toother in-vehicle IPDs (in-vehicle low-side switch ICs, in-vehicle powersupply ICs, etc.) but also to semiconductor integrated circuit devicesfor use other than in vehicles.

Furthermore, in addition to the above embodiments, it is possible to addvarious modifications to the various technical features disclosed hereinwithout departing from the spirit of the technological creation. Inother words, it should be understood that the above embodiments areexamples in all respects and are not limiting; the technological scopeof the present invention is not indicated by the above description ofthe embodiments but by the claims; and all modifications within thescope of the claims and the meaning equivalent to the claims arecovered.

INDUSTRIAL APPLICABILITY

The invention disclosed herein is applicable to in-vehicle IPDs, forexample.

LIST OF REFERENCE SIGNS

-   -   1 semiconductor integrated circuit device    -   2 ECU    -   3, 3X, 3Y load    -   4 external sense resistor    -   10, 10X, 10Y NMOSFET    -   20, 20X, 20Y output current monitor    -   21, 21′ NMOSFET    -   22 sense resistor    -   30, 30X, 30Y gate controller    -   31 gate driver    -   32 oscillator    -   33 charge pump    -   34 damper    -   35 NMOSFET    -   40, 40X, 40Y control logic section    -   50, 50X, 50Y signal input section    -   60, 60X, 60Y internal power supply    -   70, 70X, 70Y abnormality protection section    -   71, 71X, 71Y overcurrent protection circuit    -   72 open protection circuit    -   73 temperature protection circuit    -   74 voltage reduction protection circuit    -   80, 80X, 80Y output current detector    -   90, 90X, 90Y signal output section    -   91, 91X, 91Y selector    -   100 multiplexer    -   110 first current generator    -   111 operational amplifier    -   112 NMOSFET    -   113 resistor    -   120 second current generator    -   121 operational amplifier    -   122 NMOSFET    -   123 resistor    -   130, 130X, 130Y threshold voltage generator    -   131 current source    -   132 resistor    -   133 current mirror    -   140, 140X, 140Y overcurrent detector    -   141 comparator    -   150, 150X, 150Y reference voltage generator    -   151 current source    -   152 resistor    -   160, 160X, 160Y comparison section    -   161 comparator    -   170 threshold controller    -   171 comparator    -   172 current source    -   173, 173X, 173Y level shifter    -   174, 174X, 174Y RS flip-flop    -   175 discharge controller    -   176 NMOSFET    -   177 capacitor    -   178 charge controller    -   179X, 179Y delay section    -   NOR1 NOR operation unit    -   AND1 to AND3 AND operation unit    -   OR1 OR operation unit    -   INV1 to NV3 inverter    -   PG1 pulse generator    -   R1 resistor    -   C1 capacitor    -   T1 to T5, SET, DLY external terminal    -   X vehicle    -   X11 to X18 electronic apparatus

The invention claimed is:
 1. An overcurrent protection circuitcomprising: a threshold generator which switches, in accordance with athreshold control signal, whether an overcurrent detection thresholdshould be a first set value or a second set value which is lower thanthe first set value; an overcurrent detector which compares a sensesignal in accordance with a monitored current with the overcurrentdetection threshold and generates an overcurrent protection signal; areference value generator which generates a reference value inaccordance with the second set value; a comparison section whichcompares the sense signal with the reference value and generates acomparison signal; and a threshold controller which monitors thecomparison signal and generates the threshold control signal.
 2. Theovercurrent protection circuit according to claim 1, wherein when theovercurrent detection threshold has been set to the first set value, thethreshold controller generates the threshold control signal such thatthe overcurrent detection threshold is switched to the second set valueat a time point when a mask period elapses with the sense signalmaintained above the reference value.
 3. The overcurrent protectioncircuit according to claim 2, wherein when the overcurrent detectionthreshold has been set to the second set value, the threshold controllergenerates the threshold control signal such that the overcurrentdetection threshold is switched to the first set value at a time pointwhen the sense signal falls below the reference value.
 4. Theovercurrent protection circuit according to claim 2, wherein the maskperiod is a variable value.
 5. The overcurrent protection circuitaccording to claim 1, wherein the first set value is a fixed value, andthe second set value is a variable value.
 6. A semiconductor integratedcircuit device comprising, integrated therein: a power transistor whichswitches a current path, through which an output current flows, betweena conducting state and a cutoff state; an output current monitor whichgenerates a sense signal in accordance with the output current; a gatecontroller which generates a driving signal for the power transistor inaccordance with a control signal; and the overcurrent protection circuitaccording to claim 1 which monitors the sense signal and generates anovercurrent protection signal, wherein the gate controller is providedwith a function of forcibly turning off the power transistor in responseto the overcurrent protection signal.
 7. The semiconductor integratedcircuit device according to claim 6, further comprising, integratedtherein, a signal output section which selectively outputs, to outsidethe device, one of a detection result of the output current and anabnormality flag as a status notification signal.
 8. An electronicapparatus comprising: the semiconductor integrated circuit deviceaccording to claim 6; and a load connected to the semiconductorintegrated circuit device.
 9. The electronic apparatus according toclaim 8, wherein the load is a bulb lamp, a relay coil, a solenoid, alight emitting diode, or a motor.
 10. A vehicle comprising theelectronic apparatus according to claim
 8. 11. An overcurrent protectioncircuit comprising: a first threshold generator which switches, inaccordance with a first threshold control signal, whether a firstovercurrent detection threshold should be a first set value or a secondset value which is lower than the first set value; a second thresholdgenerator which switches, in accordance with a second threshold controlsignal, whether a second overcurrent detection threshold should be athird set value or a fourth set value which is lower than the third setvalue; a first overcurrent detector which compares a first sense signalin accordance with a first monitored current with the first overcurrentdetection threshold and generates a first overcurrent protection signal;a second overcurrent detector which compares a second sense signal inaccordance with a second monitored current with the second overcurrentdetection threshold and generates a second overcurrent protectionsignal; a first reference value generator which generates a firstreference value in accordance with the second set value; a secondreference value generator which generates a second reference value inaccordance with the fourth set value; a first comparison section whichcompares the first sense signal with the first reference value andgenerates a first comparison signal; a second comparison section whichcompares the second sense signal with the second reference value andgenerates a second comparison signal; and a threshold controller whichmonitors both the first comparison signal and the second comparisonsignal and generates the first threshold control signal and the secondthreshold control signal.
 12. The overcurrent protection circuitaccording to claim 11, wherein the threshold controller includes anexternal terminal for externally connecting a capacitor, a comparatorwhich compares a charge voltage which appears at the external terminalwith a predetermined reference voltage and generates an internal signal,a first flip-flop which generates the first threshold control signal inaccordance with the internal signal and the first comparison signal, asecond flip-flop which generates the second threshold control signal inaccordance with the internal signal and the second comparison signal, adischarge controller which performs discharge control of the capacitorin accordance with the internal signal, and a charge controller whichperforms charge control of the capacitor in accordance with both thefirst comparison signal and the second comparison signal.
 13. Theovercurrent protection circuit according to claim 12, wherein thedischarge controller accepts input of not only the internal signal butalso the first comparison signal, the second comparison signal, thefirst threshold control signal, and the second threshold control signal,and when, after a logic-level change occurs in one of the firstcomparison signal and the second comparison signal and a chargingoperation of the capacitor is started, a logic-level change occurs in another of the first comparison signal and the second comparison signalbefore the charge voltage becomes higher than the reference voltage, thecapacitor is discharged.
 14. The overcurrent protection circuitaccording to claim 13, wherein the threshold controller further includesa first delay section which gives a delay to the first comparison signaland generates a first delay signal, and a second delay section whichgives a delay to the second comparison signal and generates a seconddelay signal, and the first delay signal and the second delay signal,instead of the first comparison signal and the second comparison signal,are inputted to the first flip-flop and the second flip-flop,respectively.
 15. The overcurrent protection circuit according to claim11, wherein the first set value and the third set value are each a fixedvalue, and the second set value and the fourth set value are each avariable value.
 16. A semiconductor integrated circuit devicecomprising, integrated therein: a first power transistor which switchesa first current path, through which a first output current flows,between a conducting state and a cutoff state; a second power transistorwhich switches a second current path, through which a second outputcurrent flows, between a conducting state and a cutoff state; a firstoutput current monitor which generates a first sense signal inaccordance with the first output current; a second output currentmonitor which generates a second sense signal in accordance with thesecond output current; a first gate controller which generates a firstdriving signal for the first power transistor in accordance with a firstcontrol signal; a second gate controller which generates a seconddriving signal for the second power transistor in accordance with asecond control signal; and the overcurrent protection circuit accordingto claim 11 which monitors the first sense signal and the second sensesignal and generates a first overcurrent protection signal and a secondovercurrent protection signal, wherein the first gate controller and thesecond gate controller have functions of forcibly turning off the firstpower transistor and the second power transistor in accordance with thefirst overcurrent protection signal and the second overcurrentprotection signal, respectively.
 17. The semiconductor integratedcircuit device according to claim 16, further comprising, integratedtherein: a first signal output section which generates one of adetection result of the first output current and an abnormality flag asa first status notification signal; a second signal output section whichgenerates one of a detection result of the second output current and anabnormality flag as a second status notification signal; and amultiplexer which selectively outputs one of the first statusnotification signal and the second status notification signal to outsidethe device.
 18. An electronic apparatus comprising: the semiconductorintegrated circuit device according to claim 16; a first load connectedto the first power transistor; and a second load connected to the secondpower transistor.
 19. The electronic apparatus according to claim 18,wherein the first load and the second load are each a bulb lamp, a relaycoil, a solenoid, a light emitting diode, or a motor.
 20. A vehiclecomprising the electronic apparatus according to claim 18.